Amplificador - Casse D

8/7/2019 Amplificador - Casse D

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Table of Contents

1.0 Abstract .. 2

2.0 Introduction .3

3.0 Circuit Components...4

3.1 Triangle Wave Generator ..4

3.2 Pulse Width Modulator ..6

3.3 Tone control filters ..8

3.4 Volume and balance control filters ..10

3.5 H-Bridge amplification stage ..11

3.6 Demodulation filter ...12

4.0 Conclusion ....14

5.0 Calculations ......15

6.0 Circuit Schematic ..

..

18

7.0 Bill of Materials19

8.0 References and Data Sheets..20

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1.0 Abstract

This project consisted of the design and construction of a two channel, 10Watt class D

audio amplifier with a carrier frequency of 44kHz, along with volume, balance, and tone

controls. The main reason for the use of class D amplifiers is there extreme power

efficiency. Class D amplifiers are composed of a few essential components. First the

audio signal is converted into a pulse train using a pulse width modulator. That signal is

then sent through a switched mode power gain stage and then the signal is demodulated

with a low pass filter. The results of the project showed that the sound quality was

relatively good for the carrier frequency specified above. However the tone controls

were not as successful as we had hoped for especially the treble (high frequency) control.

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2.0 Introduction

Class D amplifiers use pulse width modulation techniques to achieve a very power

efficient amplifier. Class D amplifiers use transistors that are either on or off, and almost

never in-between, so they waste the least amount of power. Class B amplifiers use linear

regulating transistors to modulate output current and voltage and they can never be more

efficient than 71%. Obviously, then, class D amplifiers are more efficient than class A,

class AB, or class B. Some class D amplifiers have greater than 80% efficiency at full

power. Class D amplifiers can also have low distortion, although not as good as class AB

or class A. Because of the high power efficiency, they are ideal to use in small or

portable electronics because they do not require large heat sinks to cool the transistors.

The general components that make up the class D amplifier are a pulse width modulator,

a switched mode power gain stage, and a demodulation filter.

Class D amplifiers have been around since the 60s but were never very successful for

audio applications because they had such high distortion. With the invention of the

power MOS transistor, class D amplifiers suddenly became useful because the MOS

transistor allowed for a very fast switching frequency with little distortion. Even with the

Power MOS transistor the distortion level for high frequency signal can still be

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substantial. Today they are best used for subwoofers or low frequency signals in audio

applications.

3.0 Circuit Components

The amplifier was composed of only a few components shown in the block diagram

below.

All of the components up until the PWM will be referred to as the pre-amp in later

sections. Over the course of the descriptions and explanations to come please refer to the

complete attached schematics of the entire circuit.

3.1 Triangle Wave Generator

A triangle wave was needed to convert the audio signal into a pulse width modulated

signal using a comparator. The triangle wave generator that we made consisted of an

integrator and a hysteresis comparator. From an intuitive perspective, all the circuit did

was integrate a square wave that was created by the hysteresis comparator. The biasing

for the input and the positive feedback resistors of the hysteresis comparator were chosen

such that the output would switch to the opposite rail when the input was at +/- 10 volts.

This was done by setting the feed back resistor (R18) to 1k ohm, and analyzing the

voltage divider network between the nodes A and B. The voltage at node B is limited by

Balance

Control

Volume

Control

Tone

Controls

Pulse Width

Modulator

H-Bridge

Power AV

Demodulation

Filter Speaker

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the diodes to +/- 0.9 volts. By setting the voltage at the positive input to the op-amp to

zero(the point of witching) the value of R16 needed to switch the comparator at an input

voltage of +/- 10 volts was found to be 8.5k ohms. When the comparator flips to the

opposite rail, that signal then feeds around to the integrator, and the integrator begins to

integrate the same function but of opposite sign from before. Thus a triangle wave comes

out of the output of the op-amp at node A. If you refer to the calculations section(5.0),

greater input resistance results in a smaller slope of the output waveform and a smaller

input resistance results in a greater slope. Therefore adjusting the pot(R14) changes the

frequency of the triangle wave by dictating how fast the output of the integrator climbs

until it hits the voltage at which the comparator switches.

+VV216V

+V

V1-16V

D81N4148

D71N4148

+

U2LF351

R1410k 97%

C1.001uf

+VV416V

+V

V3-16V

+

U1LF351

R191.1k

R18

1k

R171.1k

R168.5k

R13

1.1k A B

82.8u 106u 128u 151u 174u 197u 220u-15

-10

-5

0

5

10

15

Xa: 198.3u Xb: 177.1u

Yc: 12.67 Yd:-12.00

a-b: 21.20u

c-d: 24.67

freq: 47.17k

Ref=Ground X=22.8u/Div Y=voltage 365%

d

c

b aA

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3.2 Pulse Width Modulator

The pulse width modulator took the audio signal and converted it into a pulse signal with

varying duty cycle that was proportional to the input signal at each sampled point. This

was accomplished by sending the triangle wave and the audio signal from the pre-amp

filters through a comparator. The results are shown below. As can be seen from the

oscilloscope plot of the waveforms, the triangle wave samples the input audio signal at

every point that the two curves cross and converts it into a series of pulse signals of

various lengths. A pull-up resistor was needed on the output of the comparator because it

had an open collector output. Without the pull-up resistor the output square wave had a

peak voltage of 500mv, but with the resistor the peak voltage was 16v. Converting the

audio to a PWM signal is desirable because it is better to use a square wave to drive the

Output from

pre-amp

+V

V8-16V

15kHz

V7-10/10V

U3SWCOMP

+V

V5-16V

+VV616V

+VV216V

+V

V1-16V

D81N4148

D71N4148

+

U2LF351

R1410k 95%

C1.001uf

+VV416V

+V

V3-16V

+

U1LF351

R1250

R191.1k

R18

1k

R171.1k

R168.5k

R131.1k

AB

C

PWM Signal

148u 172u 195u 219u 243u 267u 291u-18

-12

-6

0

6

12

18

Xa: 166.7u Xb: 166.7u

Yc: 15.84 Yd:-15.84

a-b: 0.000

c-d: 31.68

freq: 0.000


d

c

baA

B

C

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power gain stage because there is far less power dissipated when the transistors are turned

on and off by a square wave rather than turning them on slowly. In an effort to reduce

some of the noise and oscillations in our amplifier, we put bypass capacitors around the

positive and negative supply pins on the comparator. It helped somewhat in reducing the

noise in the circuit.

One problem that was important to avoid, and one of which we learned the hard way was

over modulation in the PWM. This occurred when the amplitude of the input signal

exceeded the amplitude of the triangle wave. When this occurred the output signal

became very distorted because there are point where the triangle wave goes through a

whole cycle without intersecting the audio signal.

357u 381u 405u 429u 452u 476u 500u-12

-8

-4

0

4

8

12

Xa: 500.0u Xb: 0.000

Yc: 10.27 Yd:-12.00

a-b: 500.0u

c-d: 22.27

freq: 2.000k


d

c

b aA

B

C

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3.3 Tone Controls

A schematic of the tone controls is shown below. The filter on the left is the filter for the

low frequencies and the filter on the right is the filter for the high frequencies. The low

frequency filter is an inverting op-amp configuration with a pot that changes the feedback

and input resistance. When the pot is turned all the way to the right the input resistance is

large and the gain is small and when it is turned all the way to the left the feedback

resistance is large and the gain is large. The capacitors are there to nullify the affect of

the pot at the frequency at which they act as a short. When that frequency is reached the

position of the pot is irrelevant and the feedback and input resistances are the same,

which yields a gain of one for all frequencies above the desired frequency. The

frequency chosen was 1kHz and can be seen in the bode plot above. So by adjusting the

1 10 100 1k 10k 100k 1Meg

-21

-14

-7

0

7

14

21

Xa: 966.7k Xb: 1.000

Yc: 20.00 Yd:-4.000

a-b: 966.7k

c-d: 24.00

Ref=Ground X=frequency(Hz) Y=(db)

d

c

b a*V(B_50

B

R12

10k .01%

C8

2.2nf

C72.2nf

+

U4LF351

+V

V8

-16V

+VV716V

+VV516V

+V

V6-16V

+

U5LF351

C21uF

C4

1uF

R110k .01%

10kHz

V2-1/1V

R41.1k

R51.1k

R6400k

R7400k

R81.11k

R91.11k

+V

V8

-16V

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pot the gain is varied between +/- 20dB for all frequencies up to 1kHz. So in reality this

acted as the bass control for the audio signal.

The high frequency filter operated in a similar but inverted fashion. Unlike the low

frequency filter where the gain was unity for all frequencies past 1kHz, the gain for the

high frequency filter is unity for all frequencies up to 1kHz. When low frequency signals

enter the filter the capacitors are open and therefore the setting on the pot has no affect on

either the feedback or input resistances. This results in a gain of one because the

feedback and input resistances are equivalent. When signals of frequencies greater than

1kHz enter the filter the pot adjust the gain of those signals by changing the ratio of

feedback resistance to input resistance. This high frequency filter was the treble control

for the audio signal. It was important to make the filters such that they did not over lap in

frequency because the two filters were connected in series. When two things are

connected in series the overall gain is the product of the gain from each individual stage.

So any signal that got amplified from one filter had a gain of unity in the other filter, so

1 10 100 1k 10k 100k 1Meg-21

-14

-7

0

7

14

21

Xa: 998.9k Xb: 1.000

Yc: 20.00 Yd: 1.000

a-b: 998.9k

c-d: 19.00

Ref=Ground X=frequency(Hz) Y=(db)

d

cb a*V(B_50

*V(B_00

*V(B_99

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that no signal would be amplified twice. The overall range of the two filters in series can

be seen in the bode plot above. The calculations for the component values can be found

in section 5.0.

3.4 Volume and Balance

I will not elaborate much on the volume and balance controls for the amplifier because

they were quite simple.

Both channels of the audio signal (left and right speakers) went through identical volume

controls. The value of the input resistance for the volume control was chosen such that

with the pot turned to 100% the volume of the music was at the desired maximum level.

The balance control was very similar. For this we wanted a gain of unity when the pot

was turned to 100%. To make the two channels work opposite to each other we used a

tandem pot with the feed back loop for the second channel connected to the opposite end

of the pot from the first channel. So when the pot was at 100% one channel had unity

gain while the other channel had a gain of zero. When the pot was at 50% the gain for

both of the channels was equivalent.

Volume Control Balance Control

+VV216V

+V

V1-16V

+

U2

LF351

100KR1

100KR2

+

U1

LF351

+V

V3-16V

+VV416V

R193.9k

R350k

R4100k

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3.5 H-Bridge

An H-bridge was used for the power amplification of the PWM signal. Shown in the

schematic below, an H-bridge is a rectangular arrangement of transistors with a load

across the center. The idea is to drive the H-Bridge with a square wave on each side of

bridge, with the driving signal on one side one half cycle out of phase from the other side.

In our circuit we used two p-channel MOSFETs for the two top transistors and three n-

channel MOSFETs, two for the bottom and one to invert the driving signal for the other

side of the bridge. We chose to use two p-channel MOSFETs for the top portion of the

bridge because we did not want to use a separate IC or have a big messy thing of circuitry

to drive a bridge made of all n-channels. P-channels turn on in exactly the opposite

manner than that of an n-channel so it made sense to drive both the gates by the same

signal. In order to make the H-Bridge work, we needed to turn on the MOSFETs in

diagonal pairs. This allowed a path of current to flow from +Vcc to ground across the

load but twice as much voltage swing because the current is in the opposite direction

across the load when the other diagonal pair turns on. By inverting the driving signal on

the right side and using that to drive the left side of the bridge, the bridge began to

function properly. The simulations showed a fairly clean amplified signal across the load

resistor with amplitude of about 8.5volts and an output power of about 12 watts. The

44.0kHz

V20/16V Q5

IRFI510G

Q4IRF9510

Q3IRFI510G

+V

V116V

Q1IRFI510G

Q2IRF9510

RL

R2100

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output power for the H-Bridge is directly proportional to the variation in duty cycle in the

PWM signal. As it was discussed earlier, when the triangle wave is much greater than

the audio signal that it is sampling the variation of the duty cycle in the PWM signal is

very small, therefor the average power dissipated across the load in the H-Bridge is very

small. When the amplitude of the input audio nears the amplitude of the triangle wave

the output power becomes much louder because the variation in the duty cycle is much

greater.

3.6 Demodulation Filter

The demodulation filter that we attempted to use was a double pole roll off LC filter. We

used this to filter the PWM signal after it has gone through the H-Bridge. The pole was

C3

.1uf

L450.0uH

10kHz

V2-1/1V

R36

A

1k 10k 100k 1Meg 10Meg-31

-24.2

-17.3

-10.5

-3.67

3.17

10

Xa: 71.59g Xb: 2.084g

Yc:-27.06 Yd:-2.982

a-b: 69.50g

c-d:-24.08

Ref=Ground X=frequency(Hz) Y=voltage(db) 199%

d

c

baA

0 17.3u 34.7u 52u 69.3u 86.6u 104u0

2.5

5

7.5

10

12.5

15

Xa: 500.0u Xb: 0.000Yc: 17.50 Yd: 0.000

a-b: 500.0uc-d: 17.50

freq: 2.000k

Ref=V(4) X=17.3u/Div Y=power 481%

d

cb a

A

0 17.3u 34.5u 51.8u 69u 86.3u 104u-9

-6

-3

0

3

6

9

Xa: 113.4u Xb: 113.4u

Yc: 8.500 Yd:-8.600

a-b: 0.000

c-d: 17.10

freq: 0.000

Ref=V(3) X=17.3u/Div Y=voltage 483%

d

c

baA

Voltage Swing Power dissipation

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placed at 20 kHz because that is the only portion of the signal that we really care about

because the human ear can only hear up to that frequency. We chose an LC filter because

it has a roll off that is twice as fast as would a first order low pass RC filter. In our actual

project, we were unable to find inductor sufficient enough to meet our needs so the Filters

were omitted due to time constraints.

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4.0 Conclusion

Overall the project was a success. This project provided a sound medium in which I

could enhance my knowledge and experience in electronics. Prior to embarking on this

project I knew nothing about class D amps, pulse width modulation, active filters, and H-

Bridges. Having spent this quarter researching and studying these things I feel I have

learned a lot and have broadened my interests and made available new subjects of interest

that I was previously unaware of. Perhaps one the most important skills that I really

developed over the quarter was trouble shooting. It is very easy to get frustrated when

things dont work out like they should, especially when it comes to soldering the final

product. Fortunately the four errors that we made in soldering up the circuit were ones

that we discover and fixed in a relatively short amount of time.

One aspect that still has me somewhat puzzled was the fact that we were never able to get

a really clean signal from our output. We tried putting bypass capacitors around the +/-

Vcc to the comparator so that it would switch with the least amount of noise in the output

wave and that seemed to give some improvement but it was not substantial. All in all I

think that this project has bettered my understanding of electrical engineering knowledge

and also time management skills.

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5.0 Calculations

Output Power

The calculated power output form the simulation is as follows.

Pout(rms) = (V2/R) , R = 6 ohms

= (8.52/6)

= 12 Wmax

Tone Controls

Treble

Case 1: Pot turned to 0%, capacitors shorted

AV = R3/(R1 + Rv//R2)


AV = (R3 + Rv//R2)/(R1)

R2R2

R1 Rv

R3

+VV412V

+V

V3-12V

+

U1LF351

R2

Rv

R2R2

R1

R3

+VV412V

+V

V3-12V

+

U1LF351

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AV = (R3 + (1/2Rv//R2))/(R1+(1/2Rv//R2))

Bass


AV = R2/(R1 + Rv)


AV = (R2 + Rv)/R1


AV = (R2 + 1/2Rv)/(R1 + 1/2Rv)

1/2Rv 1/2Rv

R2R2

R1

R3

+VV412V

+V

V3

-12V

+

U1LF351

1/2Rv

R2

RvR1

+VV412V

+V

V3-12V

+

U1LF351

R1

R2

RvR1

+VV412V

+V

V3

-12V

+

U1LF351

1/2Rv

R2

1/2RvR1

+VV412V

+V

V3-12V

+

U1

LF351

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From the results of the three gain equations for each of the filters, the resistor values were

selected and calculated to be the values labeled in the schematic.

Triangle Wave Generator

The frequency of the triangle wave is controlled by the variable resistor R 14(Refer to sect.

3.1). By changing the input resistance of the integrator, it controlled the slope of the

output waveform. This can be seen in the transfer function below.

H(s) = -1/s

Using Vin = +/- 0.9u(t), L[Vin] = +/-0.9/s

Vout(s) = -/+ 0.9/ s2

Vout(t) = -/+ (0.9/)t , = R14C1

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7.0 References And Data Sheets

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Amplificador - Casse D